RF is designed to transmit, to the Earth

RF Section Prototyping of X-band Transmitter for Remote
Sensing Satellites

Hesham A. Mohamed1, Hany B. O. Bekhit2,
and Doaa Mahmoud2

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1 Electronics Research Institute,
Giza, Egypt.

2 National Authority for Remote Sensing
and Space Sciences, Cairo, Egypt.

 

Abstract— The X-band transmitter is intended to transmit
the payload data of remote sensing satellites. The current X-band transmitter
is designed to fit in Nano and Micro satellites that require high data rate and
small budgets. The X-band transmitter carries out the functions of payload data
encoding, modulation, amplification and other functions to prepare the
satellite payload data for transmission through the satellite antenna. The
X-band transmitter is a communication subsystem, which has two main parts: the
DSP part and the RF part. The operating frequency bands ranges are (2-2.1 GHz),
and (8-8.4 GHz).

 

Index
Terms— X-band,
transmitter, remote sensing, satellite, Digital Signal Processing (DSP), Radio
Frequency (RF), payload data.

                                                                                                                                                       
I.           
INTRODUCTION

The recent advances in Earth imaging using
satellites with high resolution put high demanding to implement payload data
transmitter with high data rate to cope with satellite mission requirements.
The under development X-band equipment is designed to transmit, to the Earth
station, the group information stream at rate of up to 150 Mbps, supplied from
payload data subsystem of the satellite.

In the RF part (Fig. 1),
we intended to design, implement, manufacture, and test each module in the
design. This Paper presents the implementation of our X-band RF section design.

Fig. 1: X-band Transmitter (RF section)
Block Diagram

X-band RF section includes the following components:

·     
Frequency Synthesis
Device

·     
Power Amplifier

·     
Phase Modulator

·     
Band-Pass Filters

·     
Frequency Multiplier

·     
Low Pass Filter

·     
Up-Converter

·     
Directional Coupler

·     
Pre-Amplifier

·     
Power Detector

 

First, I/Q data streams are received from the DSP
section, then entered to our QPSK Modulator. The modulator carrier frequency is
2.05 GHz, and from the system design, the power level of the input carrier
signal should be within range (from -10 to +5 dBm), which can be generated from
a frequency synthesizer. The selected frequency synthesizer (FS) is supplied
with 10 MHz square wave signal with signal level of 1 Vp-p from the crystal
oscillator. The FS can be controlled to achieve the required frequency via
control signals from the X-band transmitter microcontroller.

The Modulator output is filtered with Band Pass
Filter and is then up-converted to the X-band (8 – 8.4 GHz) by up-converter.
The local oscillator signal to up-converter is the output from the frequency
synthesizer after multiplication by three using frequency multiplier, which
means that we have to use power splitter after the synthesizer, and because of
the selected local oscillator (LO) input of the up-converter should have power
level in range (-6 to +6 dBm), an amplifier is selected following the
multiplier to achieve the required LO signal power level.

The output signal from the up-converter is
filtered with Band Pass Filter, and then entered to the Pre-amplifier, and
because the maximum input power of the selected pre-amplifier equal to +5 dBm
and to maintain the required signal power level of the whole transmitter, we
have to use an amplifier after the up-converter. The pre-amplifier is
controlled with the Microcontroller to give the user the ability to change the
transmitter output signal level as required. The Power amplifier is used to
achieve the required signal level from the transmitter (+43 dBm).

The output signal from the power amplifier is
filtered using a low pass filter to limit the band of the transmitted signal.
The output signal power level is continuously sensed to its value as required,
this is done by passing the output signal from the amplifier to a directional
coupler and measure
the coupled signal level with a power detector.

 

This
paper will present the work done in passive circuits’ prototypes, other active
circuits under development, and integration of the RF section.

                                                                                                                                 
II. RF Microstrip circuits
(Prototypes)

In this section we design, implement, manufacture, and
test some RF microstrip circuits’ prototypes to ensure our Egyptian industries
facilities and qualifications. The implemented circuits are low-pass filter
(LPF), band-pass filter (BPF), Wilkinson power divider (WPD), and directional coupler
(DC) (?1
to ?6).

The microstrip material used in our design is FR4
and the conductive layer is copper foil.

Material specification:

Ø 
Dielectric (epsilon r)       à 4.45

Ø 
Dielectric thickness
(h)   à 1.575 mm

Ø 
Conductor thickness
(t)   à 35 µm

 

A.     
Chebyshev stepped-Impedance
L-C Ladder-Type LPF

The design of low pass filter involves two main
steps: one is to select an appropriate low pass prototype (g values),
second is the transformation of g values to L–C elements for the desired
cutoff frequency and the desired source impedance, which is normally 50 Ohms
for microstrip filters. The prototype g values for the low pass
prototype with Chebyshev filter can calculated as follows ?4:

(1)

(2)

i= 2, 3, 4,….n

(3)

(4)

(5)

Where n is the number of g elements
which can be defined as

(6)

Where

(7)

fc is the cutoff frequency

fs is the frequency at stopped
attenuation

LAS is the stopped attenuation in (dB)

LAR is the ripple value in (dB)

As a design example, choose the cutoff frequency
as 11.2 GHz and the attenuation at Fs=14.5 GHz is LAS=30
dB, while the ripple is LAR = 0.05 dB. Using the above equations,
the number of filter elements will be nine (n = 9) and the g
values for the prototype will be as given in Table 1.
Make a transformation for the g value to the low pass lumped L-C
elements based on the following equations:

(8-a)

(8-b)

So the corresponding lumped inductances and
capacitances (L-C) values will be as shown in Table-1. With represent the
inductance with high impedance transmission line (ZH=95 ? ? W2=0.36
mm), the corresponding lengths will based on the following equation:

(9)

With representing the capacitor with low impedance
transmission line (ZL = 25 ? ? W1 = 6.121 mm), the corresponding
lengths will based on the following equation:

(10)

When make the design on FR4 (?r = 4.4,
h = 1.6 mm and tan ? = 0.02), the corresponding lengths for each of inductance
and capacitance are calculated using Eqs. (9-10) and due to microstrip
tolerance and the limited ratio of (ZH/ ZL), we make
optimization for L, C values are shown in Table-1. The layout of the stepped
impedance low pass filter is shown in Fig. 2.
By ADS and CST, the simulation results for the designed low pass filter gives
(-0.245 dB) as insertion loss, and less than (-17 dB) as return loss in the
passband region. The filter achieve good stopband with cutoff frequency of 8.2
GHz and roll-off of 10.36 dB/GHz, Fig. 3.

 

Table 1: The
LPF g-values and corresponding L-C values with layout dimensions

g

L-C values after Opt.

The length (mm)

g1 = 1.0499325

L1=0.8397 nH

L1=1.6165

g2 = 1.4610869

C2=0.5447 pF

L2=2.440610

g3 = 2.0065318

L3=0.1703 nH

L3=3.581343

g4 = 1.6697470

C4=0.67 pF

L4=2.928857

g5 = 2.0857796

L5=0.1875 nH

L5=3.782665

g6 = 1.6697470

C6=0.67 pF

L4=2.928857

g7 = 2.0065318

L7=0.1703 nH

L3=3.581343

g8 = 1.4610863

C8=0.5447 pF

L2=2.440610

g9 = 1.0499320

L9=0. 8397 nH

L1=1.6165

 

Fig. 2: Chebyshev low pass filter layout.

 

Fig. 3: Simulated S-parameters of the LPF

B.     
Coupled line Band-Pass
Filter

In this section, a design of narrow band bandpass
filter is based on a parallel coupled-line bandpass structure. The response of
coupled-line filter exhibits sufficient bandwidth that covers X-band application.
The target specifications of BPF are as following: Center frequency = 8.2 GHz,
3 dB bandwidth = 400MHz (FBW = 4.07%), |S21| better than -3 dB in
the passband, |S11| less than -16 dB. Third order Chebyshev low pass
prototype filters (ripple = 0.01 dB), characteristic impedance = 50 ?, RF4 with
dielectric constant ?r =4.4, dielectric thickness h = 1.6mm,
and loss tangent = 0.02, are chosen.

Design equations for coupled microstrip lines given
in ?5
are used. The length of resonators are (l1 = l4
= 4.871 mm, and l2 = l3 = 4.68 mm). The gap
between lines are (S1 = S4 = 0.927 mm, and S2
= S3 = 2.992 mm). The width of each line (W1 =
W4 = 0.308 mm, and W1 = W4
= 0.5715 mm). The width of feed line is 2.88 mm. The layout is shown in Fig. 4.

Fig. 4: Layout of three-pole parallel
coupled-line BPF.

 

Fig. 5: Simulated and
measured transmission coefficients of the three-pole parallel coupled-line BPF.

From the simulated and measured transmission
coefficient as in Fig.5 it is found that the passband center frequency is 8.2
GHz with -3dB bandwidth of 400 MHz (4.02%).

 

C.     
Wilkinson Power Divider

The Wilkinson power divider ?1
is widely used in RF/microwave systems due to its simple circuit structure and
good isolation. In addition, broadband power divider can be easily achieved by
adopting the multi-section design.

As an example, the four-section Wilkinson power
divider in Fig. 6
is capable of achieving a greater than 10:1 bandwidth, with the useful property
of appearing lossless when the output ports are matched; that is, only
reflected power from the output ports is dissipated. The Wilkinson power
divider can be made with arbitrary power division, but we will first consider
the equal-split (3 dB) case. This divider is often made in micro-strip line or
strip-line form.

To achieve a center frequency of 2.1 GHz, and the
quarter-wave sections were the necessary 21.23 mm in length. The characteristic
impedance of the input and output transmission lines was chosen to be 50 ?. The
impedance of the quarter-wave sections was 70.7? and the connecting resistor
was 100 ?. The schematic and layout are shown in Fig. 6.

Table 2: The WPD layout dimensions

Transmission line Impedance

Line length (mm)

Line width (mm)

50

19.9093

2.94661

70.7

10.2448

1.53115

 

Fig. 6: Wilkinson
power divider layout

 

Fig. 7: Wilkinson power divider simulation
results

The simulation results for the magnitudes of S11,
S21, S22, and S23 in dB are plotted in Fig. 7.
The center frequency for each parameter is approximately 2.1 GHz, confirming
the calculations for the phase velocity in the microstrip and the quarter-wavelength
sections. Additionally, near 2.1 GHz the return losses for S11 and S22
both exceed 40 dB, or very near a reflection coefficient of zero. Similarly, S23
has a transmission coefficient that is close to zero, implying high isolation
between ports two and three. The equal-split nature of the Wilkinson is
confirmed because S21 is essentially 3 dB (50% power delivered from
port one to port two) across the band.

Typical device parameters that can be measured are
bandwidth, input and output port return losses, isolation between output ports,
and amplitude and phase balances and ripples. Manufactured Wilkinson power
dividers have been shown to achieve bandwidths. Since at least one Wilkinson
power divider was constructed by hand, the layout tolerances will be much
higher than what would be found from a manufactured power divider.
Consequently, the expected performance parameters will likely not meet the
values typical for manufactured Wilkinson power divider’s previously described.
Ideally, the measured center frequency for the port return losses should fall
within ± 100 MHz of the desired frequency in this case is 2.1 GHz.

 

D.     
Directional Coupler

Couplers are essential components for applications
in virtually all RF and microwave transmission systems, such as power and VSWR
measurements, signal sampling for monitoring or testing, equal or unequal power
division, phase shifting (particularly 90 and 180 degree), feed-forward signal
injection, isolation of signal sources. Other applications with the highest
possible performance are particularly required in instrumentation, such as the
new version of vector network analyzers require couplers with wide bandwidth,
flat frequency response, and long-term stability ?1.

It is commonly known that higher coupling in
conventional microstrip couplers can be achieved by tightening the spacing
between the coupled lines which is limited by fabrication tolerance. The
coupled microstrip lines support two propagation modes denoted as even and odd
modes.

A simple equation illustrating such a control is
given from the proportionality ?6.

where |S31| is the magnitude of the
coupling coefficient, while the ?reffe and ?reffo are the
effective dielectric constant for the even and odd modes, respectively.

Our specification:

Ø 
Center frequency         à
8.25 GHz.

Ø 
Coupling coefficient    à
30 dB

Ø 
Z0e Ohm                    à
51.6069

Ø 
Z0o Ohm                    à
48.4431

When two unshielded transmission lines are in close
proximity, power can be coupled from one line to the other due to the
interaction of the electromagnetic fields. Such lines are referred to as
coupled transmission lines, and they usually consist of three conductors in
close proximity, although more conductors can be used. The dimensions are made
variables during the optimization process as will be shown in Fig. 8.

In order to show the validity of this structure for
tight coupling and high directivity, microstrip directional coupler with
additional capacitor was designed and fabricated in case of coupling value of 3
dB and 4.7 dB. It is very difficult to achieve 3 dB coupling owing to
impractical spacing between the coupled lines in conventional edge coupled
microstrip couplers. The presented coupler was implemented by following dimension
in mm. Fig. 8
and Fig. 9
show simulation results of microstrip coupler with 3 dB and 2.7 dB coupling,
which was done by CST ?7,
respectively. Simulation results show that the designed directional coupler has
the directivity of 29 dB and 31 dB at center frequency 8.2 GHz with excellent
matched characteristic. The measured results of fabricated microstrip
directional coupler are shown in Fig. 6
and Fig. 7.
Measured results show about 3.15 dB and 4.87 dB of coupling, less 30 dB of
return loss, and 32 dB and 31 dB of directivity at center frequency 8.2 GHz,
respectively. Measured performances show excellent agreement with predicted
results.

Fig. 8: Geometry of the Directional coupler

 

Fig. 9: Simulated S-Parameters of the -20
dB initial coupler without DGS, designed at 8.25 GHz.

                                                                                                                      
III. FABRICATIONS AND
MEASUREMENT

Following the design considerations in above Sections
and simulation results, summarized in several useful design curves, the RF microstrip
circuits shown above were fabricated, using thin film technology and
photolithographic techniques. The substrate used is FR4 (?r = 4.4, h
= 1.5748 mm). The photos of the realized RF microstrip components presented
below as in Fig. 10.

After the design and simulate our RF circuits on
CST Studio Suite program and simulate we export our design as (.DXF file) and
then we create the masks that used to expose the desired part of the
photoresist.

A mask is a specialized black and white
photographic film or glass photo-plate on which a picture of the traces and
pads is printed with a laser photo-plotter.

Fig. 10: Circuits after manufacturing

 

The devices were characterized using a Vector
Network Analyzer from Agilent technologies (N9918A). A test board, using FR4
material, was designed and fabricated to connect the device under test (DUT) to
the measurement system in Fig. 11.
The SMA connectors used in the test board were previously characterized in
their own boards as in Fig. 12
in order to verify their reflection coefficients.

In order to be able to safely recover the DUT and
the test board after characterization, only the four coupler ports were
soldered to the board and the ground plane connection was made by mean of
pressure applied to the top of the device using a press system assembled for
that purpose. In that way the DUT could easily be removed after testing without
damaging the component and allowing the reusability of the test board.

Fig. 11: BPF testing

 

Fig. 12: LPF testing

                                                                                                                                   
IV. RF Circuits under
development

The X-band transmitter has RF section that consists
of some active and passive circuits. In this section we will present some
active circuits that are under development. The under development circuits are:
QPSK modulator, frequency synthesizer, frequency multiplier, upconverter,
preamplifier, and power detector.

 

A.     
QPSK Modulator

I/Q baseband inputs consist of voltage-to-current converters
that in turn drive double-balanced mixers. The outputs of these mixers are
summed and applied to an on-chip RF transformer, which converts the
differential mixer signals to a 50? single-ended output. The four balanced I
and Q baseband input ports are intended for DC coupling from a source with a
common-mode voltage level of about 0.5 V. The LO path consists of an LO buffer
with single-ended input, and precision quadrature generators that produce the
LO drive for the mixers.

Fig. 13:
QPSK Modulator (LT5528 from Linear Technology) layout

 

B.     
Frequency synthesizer and
crystal oscillator

PLL Programming can be achieved by 3-wire serial 3V CMOS
interface. The PLL (ADF4153) has a simple SPI-compatible serial interface for
writing to the device. CLK, DATA, and LE control the data transfer. When latch
enable (LE) is high, the 22 bits that are clocked into the input register on
each rising edge of SCLK are transferred to the appropriate latch. The maximum
allowable serial clock rate is 20 MHz.

Fig. 14:
Frequency
synthesizer (DSN-2620A-119+ from Mini-Circuits) layout

 

C.     
Frequency multiplier and 6
GHz amplifier

Fig. 15:
Frequency Multiplier (RMK-3-812+ from Mini-Circuits) and Amplifier (AVA-24A+
from Mini-circuits) PCB layout

D.     
Up converter

The RF port is designed for the 2 GHz to 14 GHz band and the
IF port is optimized for 500 MHz to 6 GHz operation. An integrated LO buffer
amplifier supports LO frequencies from 1 GHz to 12 GHz, requiring only 0 dBm LO
power.

Fig. 16:
Up-converter
(LTC5549 from Linear technology) PCB layout

 

E.      
Pre-amplifier

The HMC694LP4(E) is a GaAs MMIC PHEMT analog variable gain
amplifier which operates between 6 and 17 GHz. This amplifier is ideal for
microwave radio applications, where it provides up to 22 dB of gain, output P1
dB of up to +22 dBm, and up to +30 dBm of output IP3 at maximum gain, while
requiring only 175 mA from a +5 V supply. A gate bias pin (Vctrl) is
provided to allow variable gain control up to 23 dB.

Fig. 17:
Pre-Amplifier
(HMC694LP4 / 694LP4E-Hittite) PCB layout

F.      
Power detector

Fig. 18:
Power
Detector (ADL5902 from Analog Devices) PCB layout

                                                                                                                                                
V. RF System Integration

The RF section of the X-band transmitter is
composed of two boards, the modulator board and the amplifier board, where the
frequency synthesizer is integrated into the baseband module. The boards’ block
diagram are shown below. The
final boards design will be reached based on the testing results of the under
development circuits.

Fig. 19:
X-band transmitter (RF section) integration

 

Fig. 20: Modulator integrated
board layout (118.6 mm x 47 mm)

 

Fig. 21: Amplifier integrated
board layout (106 mm x 36 mm)

Acknowledgement:

This
research is funded by the STDF office Basic Projects Funds, Ministry of
Scientific Research, Egypt. Project ID: 15204. Project Name: Development of High Data Rate
X-Band Transmitter System (DTS) for LEO Remote Sensing Satellites.